Techniques for UWB communication developed from radar and other military applications, and pioneering work was carried out by Dr G. F. Ross, as described in U.S. Pat. No. 3,728,632. Ultra-wideband communications systems employ very short pulses of electromagnetic radiation (impulses) with short rise and fall times, resulting in a spectrum with a very wide bandwidth. Some systems employ direct excitation of an antenna with such a pulse which then radiates with its characteristic impulse or step response (depending upon the excitation). Such systems are referred to as carrierless or “carrier free” since the resulting rf emission lacks any well-defined carrier frequency. However other UWB systems radiate one or a few cycles of a high frequency carrier and thus it is possible to define a meaningful centre frequency and/or phase despite the large signal bandwidth. The US Federal Communications Commission (FCC) defines UWB as a —10 dB bandwidth of at least 25% of a centre (or average) frequency or a bandwidth of at least 1.5 GHz; the US DARPA definition is similar but refers to a —20 dB bandwidth. Such formal definitions are useful and clearly differentiates UWB systems from conventional narrow and wideband systems but the techniques described in this specification are not limited to systems falling within this precise definition and may be employed with similar systems employing very short pulses of electromagnetic radiation.
UWB communications systems have a number of advantages over conventional systems. Broadly speaking, the very large bandwidth facilitates very high data rate communications and since pulses of radiation are employed the average transmit power (and also power consumption) may be kept low even though the power in each pulse may be relatively large. Also, since the power in each pulse is spread over a large bandwidth the power per unit frequency may be very low indeed, allowing UWB systems to coexist with other spectrum users and, in military applications, providing a low probability of intercept. The short pulses also make UWB communications systems relatively unsusceptible to multipath effects since multiple reflections can in general be resolved. Finally UWB systems lend themselves to a substantially all digital implementation, with consequent cost savings and other advantages.
FIG. 1a shows an example of an analogue UWB transceiver 100. This comprises an transmit/receive antenna 102 with a characteristic impulse response indicated by bandpass filter (BPF) 104 (although in some instances a bandpass filter may be explicitly included), couples to a transmit/receive switch 106.
The transmit chain comprises an impulse generator 108 modulatable by a baseband transmit data input 110, and an antenna driver 112. The driver may be omitted since only a small output voltage swing is generally required. One of a number of modulation techniques may be employed, typically either OOK (on-off keying i.e. transmitting or not transmitting a pulse), M-ary amplitude shift keying (pulse amplitude modulation), or PPM (pulse position modulation i.e. dithering the pulse position). Typically the transmitted pulse has a duration of <Ins and may have a bandwidth of the order of gigahertz.
The receive chain typically comprises a low noise amplifier (LNA) and automatic gain control (AGC) stage 114 followed by a correlator or matched filter (MF) 116, matched to the received pulse shape so that it outputs an impulse when presented with rf energy having the correct (matching) pulse shape. The output of MF 116 is generally digitised by an analogue-to-digital converter (ADC) 118 and then presented to a (digital or software-based) variable gain threshold circuit 120, the output of which comprises the received data. The skilled person will understand that forward error correction (FEC) such as block error coding and other baseband processing may also be employed, but such techniques are well-known and conventional and hence these is omitted for clarity.
FIG. 1b shows one example of a carrier-based UWB transmitter 122. A similar transmitter is described in more detail in U.S. Pat. No. 6,026,125. This form of transmitter allows the UWB transmission centre frequency and bandwidth to be controlled and, because it is carrier-based, allows the use of frequency and phase as well as amplitude and position modulation. Thus, for example, QAM (quadrature amplitude modulation) or M-ary PSK (phase shift keying) may be employed.
Referring to FIG. 1b, an oscillator 124 generates a high frequency carrier which is gated by a mixer 126 which, in effect, acts as a high speed switch. A second input to the mixer is provided by an impulse generator 128, filtered by an (optional) bandpass filter 130. The amplitude of the filtered impulse determines the time for which the mixer diodes are forward biased and hence the effective pulse width and bandwidth of the UWB signal at the output of the mixer. The bandwidth of the UWB signal is similarly also determined by the bandwidth of filter 130. The centre frequency and instantaneous phase of the UWB signal is determined by oscillator 124, and may be modulated by a data input 132. An example of a transmitter with a centre frequency of 1.5 GHz and a bandwidth of 400 MHz is described in U.S. Pat. No. 6,026,125. Pulse to pulse coherency can be achieved by phase locking the impulse generator to the oscillator.
The output of mixer 126 is processed by a bandpass filter 134 to reject out-of-band frequencies and undesirable mixer products, optionally attenuated by a digitally controlled rf attenuator 136 to allow additional amplitude modulation, and then passed to a wideband power amplifier 138 such as a MMIC (monolithic microwave integrated circuit), and transmit antenna 140. The power amplifier may be gated on and off in synchrony with the impulses from generator 128, as described in U.S. Pat. No. '125, to reduce power consumption.
FIG. 1c shows a similar transmitter to that of FIG. 1b, in which like elements have like reference numerals. The transmitter of FIG. 1c is, broadly speaking, a special case of the transmitter of FIG. 1b in which the oscillator frequency has been set to zero. The output of oscillator 124 of FIG. 1b is effectively a dc level which serves to keep mixer 126 always on, so these elements are omitted (and the impulse generator or its output is modulated).
FIG. 1d shows an alternative carrier-based UWB transmitter 142, also described in U.S. Pat. No. 6,026,125. Again like elements to those of FIG. 1b are shown by like reference numerals.
In the arrangement of FIG. 1d a time gating circuit 144 gates the output of oscillator 124 under control of a timing signal 146. The pulse width of this timing signal determines the instantaneous UWB signal bandwidth. Thus the transmitted signal UWB bandwidth may be adjusted by adjusting the width of this pulse.
Ultra-wideband receivers suitable for use with the UWB transmitters of FIGS. 1b to 1d are described in U.S. Pat. No. 5,901,172. These receivers use tunnel diode-based detectors to enable single pulse detection at high speeds (several megabits per second) with reduced vulnerability to in-band interference. Broadly speaking a tunnel diode is switched between active and inactive modes, charge stored in the diode being discharged during its inactive mode. The tunnel diode acts, in effect, as a time-gated matched filter, and the correlation operation is synchronised to the incoming pulses.
FIG. 1e shows another example of a known UWB transmitter 148, described in U.S. Pat. No. 6,304,623. In FIG. 1e a pulser 150 generates an rf pulse for transmission by antenna 152 under control of a timing signal 154 provided by a precision timing generator 156, itself controlled by a stable timebase 158. A code generator 160 receives a reference clock from the timing generator and provides pseudo-random time offset commands to the timing generator for dithering the transmitter pulse positions. This has the effect of spreading and flattening the comb-like spectrum which would otherwise be produced by regular, narrow pulses (in some systems amplitude modulation may be employed for a similar effect).
FIG. 1f shows a corresponding receiver 162, also described in U.S. Pat. No. '623. This uses a similar timing generator 164, timebase 166 and code generator 168 (generating the same pseudo-random sequence), but the timebase 166 is locked to the received signal by a tracking loop filter 170. The timing signal output of timing generator 164 drives a template generator 172 which outputs a template signal and a correlator/sampler 176 and accumulator 178 samples and correlates the received signal with the template, integrating over an aperture time of the correlator to produce an output which is sampled at the end of an integration cycle by a detector 180 to determine whether a one or a zero has been received.
FIG. 1g shows a UWB transceiver 182 employing spread spectrum-type coding techniques. A transceiver of the general type is described in more detail in U.S. Pat. No. 6,400,754, to which reference may be made.
In FIG. 1g a receive antenna 184 and low noise amplifier 186 provide one input to a time-integrating correlator 188. A second input to the correlator is provided by a code sequence generator 190 which generates a spread spectrum-type code such as a Kasami code, that is a code with a high auto-correlation coefficient from a family of codes with low cross-correlation coefficients. Correlator 188 multiplies the analogue input signal by the reference code and integrates over a code sequence period and may comprise a matched filter with a plurality of phases representing different time alignments of the input signal and reference code. The correlator output is digitised by analogue-to-digital converter 192 which provides an output to a bus 194 controlled by a processor 196 with memory 198 the code sequence generator 190 is driven by a crystal oscillator driven clock 200 a transmit antenna driver 202 receives data from bus 194 which is multiplied by a code sequence from generator 190 and transmitted from transmit antenna 204. In operation coded sequences of impulse doublets are received and transmitted, in one arrangement each bit comprising a 1023-chip sequence of 10 ns chips, thus having a duration of 10 μs and providing 30 dB processing gain. Shorter spreading sequences and/or faster clocks may be employed for higher bit rates.
The transceiver described in U.S. Pat. No. 6,400,754 uses a modification of a frequency-independent current-mode shielded loop antenna (described in U.S. Pat. No. 4,506,267) comprising a flat rectangular conducting plate. This antenna is referred to as a large-current radiator (LCR) antenna and when driven by a current it radiates outwards on the surface of the plate.
FIG. 1h shows a driver circuit 206 for such an LCR transmit antenna 208. The antenna is driven by an H-bridge comprising four MOSFETs 210 controlled by left (L) and right (R) control lines 212, 214. By toggling line 214 high then low whilst maintaining line 214 low an impulse doublet (that is a pair of impulses of opposite polarity) of a first polarity is transmitted, and by toggling line 212 high then low whilst holding line 214 low an impulse doublet of opposite polarity is radiated. The antenna only radiates whilst the current through it changes, and transmits a single gaussian impulse on each transition.
FIGS. 2a to 2h show some examples of UWB waveforms, FIG. 2a shows a typical output waveform of a UWB impulse transmitter, and FIG. 1b shows the power spectrum of the waveform of FIG. 2a. FIG. 2c shows a wavelet pulse (which when shortened becomes a monocycle) such as might be radiated from one of the transmitters of FIGS. 1b to 1d. FIG. 2d shows the power spectrum of FIG. 2c. FIG. 2e shows an impulse doublet and FIG. 2f the power spectrum of the doublet of FIG. 2e. It can be seen that the spectrum of FIG. 2f comprises a comb with a spacing (in frequency) determined by the spacing (in time) of the impulses of the doublet and an overall bandwidth determined by the width of each impulse. It can also be appreciated from FIGS. 2e and 2f that dithering the pulse positions will tend to reduce the nulls of the comb spectrum. FIG. 2g shows examples of basis impulse doublet waveforms for a logic 0 and a logic 1. FIG. 2h shows an example of a TDMA UWB transmission such as might be radiated from the transceiver of FIG. 1g, in which bursts of Code Division Multiple access (CDMA)-encoded data are separated by periods of non-transmission to allow access by other devices.
Ultra wideband communications potentially offer significant advantages for wireless home networking, particularly broadband networking for audio and video entertainment devices, because of the very high data rates which are possible with UWB systems. However, UWB communications also present a number of special problems, most particularly the very low transmit power output imposed by the relevant regulatory authorities, in the US the FCC. Thus the maximum permitted power output is presently below the acceptable noise floor for unintentional emitters so that a UWB signal effectively appears merely as noise to a conventional receiver. This low power output limits the effective range of UWB communications and there is therefore a need to address this difficulty.
One way to improve the range of a UWB communications link is to adopt a rake receiver type approach to combine the energy in a plurality of multipath components of a received signal. Multipath effects arise when a signal from a transmitter to a receiver takes two or more different paths (multipaths) such as a direct path between a transmit and receive antenna and an indirect path via reflection off a surface. In a multipath environment two or more versions of a transmitted signal arrive at the receiver at different times. Most wireless environments, and in particular indoor environments, have significant levels of multipath which, in a conventional RF communications system, typically produces a comb-like frequency response, the multiple delays of the multipath components of the received signal giving the appearance of tines of a rake. The number and position of multipath channels generally changes over time, particularly when one or both of the transmitter and receiver is moving.
It is helpful to briefly review the operation of a conventional rake receiver before going on to consider a known UWB rake-type receiver.
In a spread spectrum communication system a baseband signal is spread by mixing it with a pseudorandom spreading sequence of a much higher bit rate (referred to as the chip rate) before modulating the rf carrier. At the receiver the baseband signal is recovered by feeding the received signal and the pseudorandom spreading sequence into a correlator and allowing one to slip past the other until a lock is obtained. Once code lock has been obtained, it is maintained by means of a code tracking loop such as an early-late tracking loop which detects when the input signal is early or late with respect to the spreading sequence and compensates for the change. Alternatively a matched filter may be employed for despreading and synchronisation.
Such a system is described as code division multiplexed as the baseband signal can only be recovered if the initial pseudorandom spreading sequence is known. A spread spectrum communication system allows many transmitters with different spreading sequences all to use the same part of the rf spectrum, a receiver “tuning” to the desired signal by selecting the appropriate spreading sequence (CDMA—code division multiple access).
One advantage of conventional spread spectrum systems is that they are relatively insensitive to multipath fading. A correlator in a spread spectrum receiver will tend to lock onto one of the multipath components, normally the direct signal which is the strongest. However a plurality of correlator may be provided to allow the spread spectrum receiver to lock onto a corresponding plurality of separate multipath components of the received signal. Such a spread spectrum receiver is known as a rake receiver and the elements of the receiver comprising the correlator are often referred to as “fingers” of the rake receiver. The separate outputs from each finger of the rake receiver are combined to provide an improved signal to noise ratio (or bit error rate) generally either by weighting each output equally or by estimating weights which maximise the signal to noise ratio of the combined output (“Maximal Ratio Combining”—MRC).
FIG. 3a shows the main components of a typical rake receiver 300, A bank of correlators 302 comprises, in this example, three correlators 302, 302 and 302 each of which receives a CDMA spread spectrum signal from input 304. The correlators are known as the fingers of the rake; in the illustrated example the rake has three fingers. The CDMA signal may be at baseband or at IF (Intermediate Frequency). Each correlator locks to a separate multipath component which is delayed by at least one chip with respect to the other multipath components. More or fewer correlators can be provided according to a quality-cost/complexity trade off. The despread output from a correlator is a signal with a magnitude and phase modified by the attenuation and phase shift of the multipath channel through which the multipath component locked onto by the finger of the rake receiver has been transmitted. A channel estimate comprising a complex number characterising the phase and attenuation of the communications channel, in particular for the multipath component of the channel the rake finger has despread, may be obtained, for example using a training sequence. The channel estimate may then be conjugated to invert the phase (and optionally normalised) and used to multiply the received signal to compensate for the channel.
The outputs of all the correlators go to a combiner 306 such as an MRC combiner, which adds the outputs in a weighted sum, generally giving greater weight to the stronger signals. The weighting may be determined based upon signal strength before or after correlation, according to conventional algorithms. The combined signal is then fed to a discriminator 308 which makes a decision as to whether a bit is a 1 or a 0 and provides a baseband output. The discriminator may include additional filtering, integration or other processing. The rake receiver may be implemented in either hardware or software or a mixture of both.
The effects of multipath propagation on UWB transmissions are not the same as on conventional RF transmissions. In particular where a UWB signal comprises a succession of wavelets or pulses (the terms are used substantially synonymously in the specification), because of the short duration and relatively long separation (in time) of these pulses it is often possible to substantially time-resolve the pulses belonging to multipath components of the UWB signal. In simple terms, the delays between the arrival of pulses in different multipath components originating from a single transmitted UWB pulse are often long enough to make it unlikely that two pulses arrive at the same time. This is described further below and can be exploited to advantage in a UWB receiver design.
It is known to apply conventional rake receiver techniques to UWB communications systems, as described for example in WO01/93441, WO01/93442, and WO01/93482. FIG. 3b, which is taken from WO01/93482, shows such a transceiver; similar arrangements are described in the other two specifications.
Referring to FIG. 3b, this shows a UWB transmitter 70, 21, 17, 23, 25, 27, 1 and a UWB receiver 1, 27, 3, 29, 31, 1-N, 7 1-N, 9. The receiver comprises a plurality of tracking correlators 311-31N together with a plurality of timing generators 71-7N, and as described in WO '482 (page 15) during a receive mode of operation the multiple arms can resolve and lock onto different multipath components of a signal. By coherent addition of the energy from these different multipath signal components the received signal to noise ratio may be improved. However the design of '482 is relatively physically large, expensive and power hungry to implement and fails to take advantage of some aspects of UWB multipath transmission.